Making my 32kHz Crystal Oscillator Actually Oscillate

For Project Whack a Mole I need a 32.768kHz crystal oscillator. I found this circuits on the Interwebs and gave it a try:

It wouldn’t go. I messed about changing component values for while, then decided to actually try to understand the circuit. Now for an oscillator to work, we need an amplifier with a gain of greater than 1, and a phase shift of 360 degrees to get positive feedback.

The circuit above is an amplifier, with the crystal network connected between the collector output and base input. We get half of the 360 degree phase shift by using a common emitter topology, which is an inverting amplifier. So the crystal network must provide the other 180 degrees. On a good day. If it’s working.

First problem – the transistor was saturated, with Vc stuck near 0V. For an oscillator to start noise gets amplified, filtered by the crystal, amplified again etc. I reasoned that if the amplifier wasn’t biased to be linear, the oscillations couldn’t build up. So I reduced the collector resistor to 6.8k, and changed the the base bias resistor to 1.8M to get the collector voltage into a linear region. So now we have Vc=3.2V with a 5V supply.

But it still wouldn’t go. On a whim I adjusted the supply voltage up and then down and found it would start with a supply voltage beneath 3V, but not any higher. Huh?

Much fiddling with pencil and paper followed. Time for a LT Spice simulation of the “AC model” of the circuit:

I’ve “opened the loop”, to model the collector driving the crystal network which then drives the base impedance. On the left is a voltage source and 6.8k resistor that represents the collector driving the 330k resistor and an equivalent model of the crystal.

The values Lm, Cm, Rm, are the “motional” parameters. They are what the mechanical properties of the crystal look like to this circuit. The values are amazing, unrealizable if you are used to regular electronic parts. I found Cm = 1fF (1E-15 Farads, or 0.001 pF) in a 32kHz crystal data sheet, then solved f=1/(2*pi*sqrt(LC)) for Lm to get the remarkable value of 24,000 Henrys. Wow.

Phase Shift

With Vcc=5V, we have Vc=3.2V, so a collector current Ic = (5-3.2)/6800 = 0.265mA. I’m using a small signal transistor model, with the emitter resistance re=26/Ic = 26/0.265 = 100 ohms. The effective impedance looking into the base rb=beta*re = 100*100 = 10k ohms (2N3904’s have a minimum beta of 100).

OK, so here is the phase response near 32kHz:

Well it looks about right, a phase shift of 170 degrees, which is close to the target of 180 degrees.

Now, can we explain why the oscillator starts with a reduced supply voltage? Well, reducing Vcc would reduce Ic and hence increase rb, the base impedance the crystal network is driving. So lets double rb to 20k and see what happens to the phase:

It gets closer to 180 degrees! Wow, that means it is more likely to oscillate. Just like the actual circuit.

So – can I induce it to oscillate on a 5V supply? Setting rb back to 10k, I messed about with C1 and C2. Increasing them to 82pF moved the phase shift to just on 180 degrees. I soldered 82pF capacitors into the circuit and it started on a 5V rail. Yayyyyy. Go Spice simulations.

Loop Gain

But what about the loop gain? Well here is the magnitude plot near 32kHz:

The maximum gain is -22dB at series resonance, followed by a minimum gain at parallel resonance. We need a net gain around the loop of 1 or 0dB. So the gain of the amplifier must be at least +22dB to get a net gain of 0dB around the loop.

A net gain of 0dB is enough to sustain oscillation, but to get it started we need a gain of greater than 0dB to amplify internal noise up to the point where we have a useful output voltage. This paper suggests a gain margin of 5 or 14dB.

The gain of a common emitter amplifier is Rc/re = 20*log10(6800/100) = 36dB, which is just the 14dB gain margin we need. At the reduced supply voltage lets say Ic is halved, so re doubles. This would reduce the loop gain to 30dB. However rb=beta*re would also double to 20k. Spice tells me the maximum gain of the crystal network is now -16dB, as rb=20k loads the circuit less. So once again we have our 14dB gain margin, which predicts the oscillator will start – which is what happens in the real hardware.

Increasing C1 and C2 to 82pF produced a crystal network gain of -24dB. With a 5V supply the amplifier gain is 36dB so we have a little less margin (12db) than we would like, but still close enough and well above 0dB. It takes about 10 seconds for the oscillations on the collector to hit the supply rails.

Start Up Time

I did some reading on this. At start up, we can model the oscillator as as a noise source being band pass filtered by the crystal, then amplified. This is then fed back to the input of the circuit and the cycles repeats, the “band pass noise” getting larger every time.

It’s humbling to think that our magically stable, low phase noise crystal oscillators are really just band pass noise that has been amplified. An oscillator is a narrow band noise source.

Every resistor generates thermal noise. The biggest resistor I can see is the 330k in series with the input. A useful rule of thumb is every 50 ohm generates 1nVrms of noise per 1 Hz of bandwidth at room temperature. So that’s our initial noise source. It’s value probably affects start up time.

OK, so what is the bandwidth (BW) of the crystal “band pass filter”. Well for a resonant circuit Q = f/BW = Xl/R. With a little manipulation and plugging the crystal motional parameters I get BW = Rm/(2*pi*Lm) = 0.225 Hz. That’s pretty narrow, which is what we would expect from a crystal I guess.

The bandwidth of a filter affects it’s delay. It takes some time for the band pass noise energy to pass through the crystal, get amplified, then be fed back once again for another lap. That sounds like exponential growth to me. We can describe the delay in terms of the filter time constant, Tau. Given the bandwidth BW we can find Tau = 1/(2*pi*BW) = 0.707 seconds. I suspect Tau would be affected by the filter shape factor so it’s an approximation for the crystal BPF. But engineers like approximations, as long as the rockets don’t blow up and bridges don’t fall down.

So we start with noise from (mainly) the 330k resistor in a 0.225Hz bandwidth. If 50 ohms gives us 1nV in 1Hz bandwidth, then 330k gives us an initial input voltage V1 = (330E3/50)*0.225*1E-9 = 1.48uVrms. Lets say the final voltage is V2 = 1Vrms (2.8Vpp) before the amplifier starts to clip and it settles down to a steady state output voltage.

The voltage grows exponentially from the initial resistor noise V1 to the final voltage V2. Plugging this into a formula for exponential growth we have V2 = V1*g*exp(t/Tau), where g is the voltage gain of 5 (14dB), and t is the start up time. Messing with logs I get t = (ln(V2) – ln(V1) – ln(g))*Tau = 8.3s

Whooo! Which is about the start up time of the real circuit.

Before I couldn’t even speel Ingineer. Now I are one.

Crystal Power

Matt, VK5ZM, suggested the function of the 330k resistor is to limit the power through the crystal. These tiny crystals are rated at just 1uW maximum power. With 1Vrms AC drive, Spice measured a current of 7.2uA through the crystal series resistance Rm=35k at the resonant frequency, which is a power of 35E3*(7.2E-6)^2 = 1.8uW. Oops, a bit much. However I think increasing the 330k resistor might reduce the loop gain. And I have a big bag of spare crystals.

Matt, and Erich, VK5HSE also pointed out there are some parasitic capacitors from the transistor that should be included in the model. The values for these capacitors is difficult to determine. My best guess from the 2N3904 data sheet and reading about transistors is Ccb=4pF (between collector and base), and Cbe=15pF (between base and emitter). Cbe could be absorbed into C1 which will add a little more phase shift, perhaps explaining why my phase plots above are just shy of 180 degrees. Ccb would be across the entire crystal network. It’s impedance at 32kHz is Xc=1/(2*pi*32E3*4E-12)=1.2M so it probably doesn’t have much effect.

Here is the final circuit that works on 5V:

John’s Solution

John has suggested the original circuit may have a wiring error. He used the circuit at the top of this post (R1=3M3, R2=68k), but connecting R1 between the base and collector, rather than base and Vcc. See Johns comments below.

Links

Open Loop LT Spice simulation of the crystal oscillator network.

10 thoughts on “Making my 32kHz Crystal Oscillator Actually Oscillate”

  1. Hi All,

    Another GREAT analysis :). I have never been a believer in simulations….

    I guess I should reconsider !!

    Another side effect of the 330k resistor is to make the shape of the output much more square which can also be very useful.

    Lots of fun :).

    John

  2. Hi All,

    As I was looking at the final circuit, it occurs to me that if the 1.8 meg resistor is returned to the collector rather than the supply, the transistor will adjust itself to its linear region regardless of some beta variation. Was the original circuit miscopied?

    John

    1. Yes the original is correct – I found the same circuit on a couple of sites by Googling “32kHz clock oscillator”.

      The DC bias arrangement you suggest would be more stable. Some care would need to be taken to ensure it doesn’t affect the AC response (loop gain and phase), given the high impedance of the crystal network.

    1. Reasonable? Hee. Well, if you want a DC bias in a 3 Volt circuit, with a 1 Volt signal. As I read though, it seems the author was looking for a 5 volt circuit with enough drive to feed a mixer.

      1. Well valid points, but that was not in my primary focus. I was more after stabilisation, i.e. Re, bias of Base etc.

        Of course the output levels, as well as the lack of a series resistor leave enough room forsome serious engineering fun!

  3. Hi All,

    That circuit looks more reasonable. However, the remaining question is crystal overdrive. Also, the waveshape at the collector will not be very square.

    So I guess crystal drive level of this circuit is a question for David and his simulator.

    I should probably try the original circuit with the hi meg resistor returned to the collector instead of the power rail.

    Lots of fun :).

    John

  4. Hi David,

    Earlier today I built the original circuit except that I returned the 3.3 meg resistor to the collector of the 2n3904.

    Without the crystal present the collector of the 2n3904 was 2.49 volts.

    With the crystal [very small!] in the circuit it takes about 3 seconds to come up to full amplitude and the collector waveshape is pretty square and the fall time is pretty fast.

    So, for what it is worth, I am convinced that the ‘original’ of the circuit given had the 3.3 meg resistor going to the collector of the 2n3904.

    Lots of fun :).

    John

    1. Nice work John!

      So your bias arrangement brought the circuit into linear territory. Rc is higher and rb would be much higher due to reduced Ic. This would load the crystal network less, leading to higher loop gain and faster start up. Phase shift will also change but we would need to simulate to work out the effects of that.

      You may have solved the mystery…..

  5. Hi David,

    This ‘chapter’ would make a good addition to some electronics book.

    Maybe what you ‘know’ is true isn’t….

    Haven’t we all learned that :).

    John

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